反激的典型波形讲解

1、反激变换器的例子Analysisof basicwaveforms基本波形分析The analysisof the basic waveformswill be done on a simula ted exampleof a flyback converteroperating in discontinuouscond uction mode.Typical drain-sourcevoltage waveform of the primary side switch is shown in Fig. 16.在电感电流断续模式下运行的反激变换器的典型一次侧 漏源极开关电压波形见图1 6 o&

2、gt;善s与30Fig. 16 Typical drain-sourcevoltage of the MOSFET in a flyback图1 6反激变换器的典型漏源极电压Thesedrain-sourcevoltage waveformscan be theoretically distin guishedinto typical elementsDifferent physical phenomenanfluen ce the waveform at given time interval. Fig. 17 and Tab. 4 demo nstrate the main element

3、sof the voltage waveform. The superposit ion of all theseelementsresults in a typical drain-sourcevoltage s hown in Fig. 16.这些漏源极电压波形能用典型的理论来描述。各个时间段有不同物理现象影响这些波形。图1 7和平台4描述了电压 波形的主要原理。把这些原理按时序整合呈现由图1 6所示 的典型漏源极电压。Fig. 17 Main elementsof the drain-sourcevoltage图1 7漏源极电压的主要原理Element 1:voltage fall du

4、ring turning on原理1 :开通期间的电压下降过程Element 2:parasitic oscillation during turning on due to current spike2011-io-20105 10原理2:在开通期间因寄生震荡产生的电流尖刺Element 3:voltage rise during turning off原理3 :关断期间的电压上升Element 4: clamping voltage of snubberso06040205.6 10 '5.S -10原理4:缓冲电路的钳位电压Element 5:parasitic oscillat

5、ion after clamping involving mainly the output capacitance of the MOSFET and the leakage inductance of the transformer原理5:钳位过程结束后主要由场效应晶体管输由电容和变压器漏感引起的寄生振荡500-502.S3 io-6 3.2 io-*1-100Element 6:parasitic oscillation after flyback phase involving mainly the output capacitance of the MOSFET and the ma

6、gnetization inductance of the transformer100原理6:磁芯存储磁能释放完毕后主要由场效应晶体管输由电容和变压器电感引起的寄生振荡EiemenlZ：reflected voltage during Hie fl/back phaseElement a：main rectangular signal with bus ampfitude, $府"夕 msuf ： i*原理7:反激变换器释放磁能期间的反射电压原理8:与直流母线电压等幅的主要方波Tab. 4 Main elementsof the drain-sourcevoltage平台4漏源极电

7、压的主要原理The spectrumof the whole drain-sourcewaveform (Fig. 16) is presentedn Fig. 18.图1 6所示的漏源极电压呈现的电磁干扰频谱见图1 8 ofrequency. Hz9SFFig. 18 Spectrumof the drain-sourcevoltage (as shown in Fig. 16)图1 8 图1 6所示的漏源极电压呈现的电磁干扰频谱The spectraof the main elementsof the drain-sourcevoltage c an be found in Fig. 20

8、. Fig. 19 is exactly the sameas Fig. 17 and has been repeatedhere for better under-standing.图2。描述了漏源极电压主要原理产生的电磁干扰频谱。 为便于理解，将图1 7映射成图1 9。Fig. 19 Main elementsof the drain-sourcevoltage (repeatedsameas Fig. 17)图1 9漏源极电压的主要原理(正确重复 图1 7 )Fig. 20 Spectraof the main elementsof the drain-sourcevoltage图2 0

9、漏源极电压主要原理产生的电磁干扰频谱This method allows associatingcertain parts of the spectrum with their root causesj.e. the peak at 20 MHz in the spectrum of the drain-sourcevoltage is causedby the parasitic oscillation d ue to the output capacitanceof the MOSFET and the leakagein ductanceof the transformer.这种方法可以

10、确定电磁干扰频谱中某些频点的来源，也就是说漏源极电压产生的电磁干扰频谱中的2。兆赫兹峰点 是钳位过程结束后主要由场效应晶体管输由电容和变压器 漏感引起的寄生振荡产生的。The analysisof the drain current of the primary switch will b e donein the sameway. Fig. 21 demonstrates typical drain cur rent in a DCM flyback.对一次侧开关的漏极电流进行分析采用相同的方法。图21展示由一个工作于电感电流断续模式反激变换器的典型漏极电流。Fig. 21 Typical

11、drain current in a flyback图2 1反激变换器的典型漏极电流This waveform can be presentedas a superpositionof the following elements(Fig. 22 and Tab. 5). The superpositionof all theseelementsresults in a typical drain current shown in Fig. 21.这个波形可以被看作是下列原理的叠加(图2 2和平台5) o全部这些波形的叠加整合结果变成图2 1所示的典型 漏极电流。amWi i ssT®

12、; trrrw £. JiHflP13PFig. 22 Main elementsof the drain current图2 2漏极电流的主要原理Element V main trianale of the drain current原理1 ：漏极电流的主要三角波形Element 2:current spike during turning on due to parasitic capacitances of the circuit原理2:在开关开通期间因寄生分布电容引起的电流尖刺Element 3:parasitic oscillation after clamping inv

13、olving mainly the output capacitance of the MOSFET and the leakage inductance of the transformer原理3:钳位过程结束后主要由场效应晶体管输由电容和变压器漏感引起的寄生振荡Element 4: parasitic oscillation after flyback phase involving mainly the output capacitance of the MOSFET and the原理4:磁芯存储磁能释放完毕后主要由场效应晶体管输由电容和变压器电感引起的寄生振荡Tab. 5 Main

14、elementsof the drain current平台5漏极电流的主要原理The spectrumof the whole drain current waveform (Fig. 21) is presentedn Fig. 23.全部漏极电流波形产生的电磁干扰频谱（图2 1 ）呈现在图2 3。Fig. 23 Spectrumof the drain current (as shown in Fig. 22)flIDw 肿 口 4图2 3漏极电流产生的电磁干扰频谱(与图2 2相同)The spectraof the main elementsof the drain current c

15、an be found in Fig. 25. Fig. 24 is exactly the sameas Fig. 22 and ha s beenrepeatedfor better understanding.漏极电流主要原理产生的电磁干扰频谱见图2 5。图2 4和图2 2相同i hT*i.s -w-11 tiimp £ ?-ifl_*h i.Fig. 24 Main elementsof the drain current图2 4漏极电流的主要原理Fig. 25 Spectraof the main elementsof the drain current 图2 5漏极电流主

16、要原理产生的电磁干扰频谱As in caseof drain-sourcevoltage this methodallows to associ ate the elementsof the drain current waveform with its contributi on to the whole spectrum.For example,the peak at 20 MHz inthe spectrumis causedby the parasitic oscillation due to the out put capacitanceof the MOSFET and the l

17、eakageinductanceof t he transformer.就象漏源极电压的例子那样，用这种方法也可以找由漏极电流的哪一部分对电磁干扰频谱产生影响。举例说明，2 0 兆赫兹的峰点是钳位过程结束后主要由场效应晶体管输由 电容和变压器漏感引起的寄生振荡产生的。This methodof separatingthe waveform in time domain into i ts main elementshelps to find out what part of the spectrumin frequencydomain causedby what related physica

18、lphenomenaTh e separationinto main elementsshould be done in respectof reaso nable eventsin the power circuit like on and off slopes,oscillatio ns, clamping,snubbering,reflectedvoltage, etc.这种在时域里对主要原理进行拆分的方法有助于我生产 生电磁干扰频段的干扰源。这种离析主要原理的手法有助于 合理审视电源电路里诸如变化速率、振荡、钳位、缓冲、反 射电压等过程。In this flyback exampleo

19、nly the primary switch has beenanal yzed as active sourceof electrical noise.There are also others,like secondaryside diodes or synchronousectifier, control IC (especiall y its gate drive), etc. In order to obtain morecompleteanalysisal l theseinterferencesourceshave to be analyzed.在这个反激变换器里只对一次侧开关进

20、行电磁噪声产生 的分析。但是还有其他的部分，象二次侧的二极管或同步整 流器、控制集成电路（尤其是它们的栅极驱动）等等。按顺 序分析将获得更完善的关于这些电磁干扰源的解析。However,it is impossibleto predict the conductedEMI spectr um using this approachdue to the fact, that only interferencesou rces are consideredThereis no analysis of the spreadingpaths of the interferencein this met

21、hod.然而，这种方法不可能预知用频谱反映的电磁干扰的实际 行为，仅仅是干扰源被重视起来。在那里没有对分布参数产 生的干扰进行分析的方法。Neverthelessthe associationof harmonicsroot causewith the respectecphysical phenomenavill reducethe efforts of EMI reduc tion. The impact of the identified root causecan be reducednot only by filtering, but also by meansof influenc

22、ing the root cause itself.不过，重视物理现象并不能成就电磁干扰的降低。降低干扰并不仅仅是滤波，也同样意味着干扰源自身的影响。Operationmodesof discontinuouslyback converter电感电流断续工作反激式变换器的运行模式The flyback converterrunning in discontinuousconductionmode can be operatedin hard switching or quasi resonant (or valley switching, or ZVS) moderegardingthe pr

23、imary side switch. The differencebetween a hard switching and quasi resonantflyback co nverter is the turn on time point of the primary switch. In a har d switching mode the turning on of the MOSFET is not synchro nized with the drain-sourcevoltage value. This type of converters runs mainly in fixed

24、 frequencymode.电感电流断续工作的反激式变换器一次侧开关可工作于 硬开关或准谐振(或谷值开关或零电压开关)模式。硬开关 和准谐振反激变换器之间的差异在于一次侧开关的开启时 间点。在硬开关里场效应晶体管的开启波形拐点并不和漏源 极电压值同步。这种变换器大体上运行于固定频率模式。In a quasi resonantmodethe resonantcircuit determinedby t he output capacity of the MOSFET and the inductanceof the tr ansformerwill be utilized to switch

25、on at lowest possiblevalue o f the drain-sourcevoltage. This circuit starts to oscillate at the en d of the current flow through the secondaryside of the transform er, henceat the end of the flyback phase.The MOSFET will be turned on at the minimum of this oscillation. The quasi resonant approachuse

26、sthis oscillation to achieveminimum voltage switching during turn on for the MOSFET. This operation moderuns at a variable frequency.在准谐振模式里，由变压器电感和场效应晶体管输由电容 引起的谐振促使开关的开通时刻发生在漏源极电压的最小 值上。这种电路在电流从变压器二次侧流尽以后（反激回扫 过程结束）开始振荡。场效应晶体管将在振荡幅值的最小值 开启（谷值开通）。这种运行模式工作在可变的频率上。Higher amplitude of the oscillation

27、results in lower drain sourc e voltage level at which the MOSFET turns on correspondingly wer switching lossesand higher efficiency of the system.更高幅值的振荡导致场效应晶体管更低的漏源极开通电压幅值来产生更低的开关损耗和更高的系统效率。To achievehigh oscillation peaks,the designof the transformer has to be set to high reflectedvoltage. This i

28、ncreaseof the refle cted voltage results in a higher drain-sourcevoltage blocking MOS FET and longer duty cycles.要达到比较高的振荡电压峰值，变压器的反射电压必须设置的比较高。增加的反射电压导致使用更高漏源极击穿电压 的场效应晶体管和更大的开关占空比。Comparisonof three different flyback solutions has beenmade. All of them have beenoperation at 300 kHz, bus voltage of

29、 40 0 V, output power of 120 W, output voltage of 16 V. Thesede sign included different modesof operation and different values of reflectedvoltage, resultng in different MOSFET s voltage ratings:比较现有的三种反激变换器。它们都工作在3 0。千赫兹，直流母线电压4 0 0伏特，输由功率1 2 0瓦特，输由电压1 6伏特。这些设计包含不同的运行模式和反射电压等级，因此使用不同电压等级的场效应晶体管：Har

30、d switching flyback with CoolMOS 600V, reflectedvoltage of 100V硬开关反激变换器使用6 0 0伏特 CoolMOS 1 0 0伏 特反射电压Quasi resonantflyback with CoolMOS600V, reflectedvoltage of 100V准谐振反激变换器使用6 0 0伏特 CoolMOS 1 0 0伏 特反射电压Quasi resonantflyback with CoolMOS800V, reflectedvoltage of 390V准谐振反激变换器使用80 0伏特CoolMOS 390伏特反射电压

31、The clampingsnubbercircuit was set to the rated breakdownvoltage of the MOSFET (600 V and 800 V respectively).钳位缓冲电路被设定在场效应晶体管的额定击穿电压上（分别为6 0 0伏特和8 0 0伏特）Flyback in hard switching modewith 600V MOSFET 使用6 0 0伏特场效应晶体管的硬开关反激变换器The hard switching approach(as shown in Fig. 26) doesn 'ct onsiderthe m

32、inimum drain-sourcevoltage. The MOSFET will be t urned on hard, in this caseat a voltage level of 500 V (at time point 3.3 以 s)The dischargeof circuits parasitic capacitancesea ds to a high current spike during turning on.硬开关(图2 6所示)几乎不考虑漏源极电压的最小值。 场效应晶体管开通应力大，在这个例子里，开通电压在5 0 。伏特(在3 .3微秒的时间点)。由寄生电容引

33、起的泄放电 流在开通时产生很高的电流尖刺。Fig. 26 Drain-sourcevoltage and drain current of hard switching600V flyback图2 66 0 0伏特硬开关反激变换器的漏源极电压和漏极电流Flyback in quasi resonantmodewith 600 V MOSFET 使用6 0 0伏特场效应晶体管的准谐振反激变换器The drain-sourcevoltage (Fig. 27) starts oscillating at the end of the flyback phaseand reachingthe min

34、imum of 300 V when the MOSFET turns on.漏源极电压(图2 7)在反射过程结束后并减小到3 0 0 伏特时场效应晶体管导通。The duty cycle is lower comparedto an 800 V solution due to a lower reflectedvoltage of 100V. Shorter duty cycle for the sameoutput power results in higher peak currents on the primar y side.因为1 0。伏特的反射电压，比较8 0 0伏特解决方案 它

35、有更小的占空比。小占空比实现同样的功率输生必须使用 更高的一次侧峰值电流。Fig. 27 Drain-sourcevoltage and drain current of quasi resonant600V flyback图2 76 0 0伏特准谐振反激变换器的漏源极电压和漏极电流Flyback in quasi resonantmodewith 800 V MOSFET 使用8 0 0伏特场效应晶体管的准谐振反激变换器The drain-sourcevoltage (Fig. 28) starts oscillating at the end of the flyback phaseand

36、 reachingthe minimum of 100V when t he MOSFET turns on. The turning on current spike is low.漏源极电压(图2 8)在反射过程结束后并减小到1 0 0 伏特时场效应晶体管导通。开通电流尖刺比较低。The duty cycle is higher comparedo a 600V solution due t o a higher reflectedvoltage of 390V. Longer duty cycle for the s ame output power results in lower p

37、eak currents on the primary s ide.因为有3 9。伏特的反射电压，所以有比6 0 0伏特解 决方案更大的占空比。更大的占空比实现同样的输生功率可 以使用更低的一次侧峰值电流。O.E4OO l,E-06Fig. 28 Drain-sourcevoltage and drain current of quasi resonant800V flyback图2 88 0 0伏特准谐振反激变换器的漏源极电压和漏极电流Comparisoof spectra干扰频谱比较The spectraof the drain-sourcevoltagesfor correspondinglyba ck design(Fig. 26Fig. 27 and Fig. 28) are shown in Fig. 29.相应设计的反激变换器(图2 6、图2 7和图2 8)的漏源极电