LTCC带通滤波器外文翻译.doc

来源：Microwave Conference,2008.EuMc 2008.38th EuropeanBandpass Filters for Ka-Band Satellite Communication Applications Based on LTCC Abstract -The design of two compact bandpass filters for Ka-band satellite communication applications (downlink 1722 GHz) is presented. Both filters are designed with additional transmissionzeros at finite frequencies in order to improve the out-of-band selectivity. The filters have been realised as low-cost LTCC modules and the scattering parameters have been measured with on-wafer probing.I. INTRODUCTION Modern multimedia satellites are getting more complex with transponder connectivity requirements, multiple coverage beams, et cetera. Hence, satellite operators increasingly demand flexibility in function, efficient signal routing and signal processing. Therefore, future applications in multimedia satellite communications require innovative components with high RF-performance and, at the same time, with low weight, small size, and high reliability. Advanced integration and packaging technologies, such as low temperature co-fired ceramics (LTCC), combine different design techniques with low cost of fabrication, small size and multiple functionality. LTCC technology provides modular,hybrid-integrated systems with a high degree of miniaturization of microwave payload equipment and, hence, flexibility for adaptation to varying applications. It is one of the reasons,why LTCC became very popular, not only for low frequencydesigns, but for high frequencies in the microwave and even millimeter-wave ranges as well . This paper focuses on bandpass filters for Ka-band downlink frequencies as important parts of multimedia satellite signal chains, which are based on LTCC multilayer technology and can be combined with other Ka-band microwave LTCC modules presently under development. The two filters presented here display high stop-band isolation for an efficient suppression of the Ka-band uplink frequencies, and low insertion loss. They are developed for separating channel groups with total bandwidths of 500 MHz up to 1 GHz in the frequency range of 1722 GHz. For demonstration purposes, these filters have been developed for a centre frequency of f0 = 19.5 GHz. Additional effort has been spent in investigating the influence of grounding and different conductive pastes on the filter performance.II. DESIGN OF LTCC BANDPASS FILTERS WITH TRANSMISSION ZEROS AT FINITE FREQUENCIES The following specifications have been chosen for the design of the bandpass filter: -Passband: 18000 21000 MHz; -Maximum in-band insertion loss: 2 dB; -Minimum in-band return loss: 12 dB (VSWR 1.7); -Steepness of the filter slopes: 20 dB/GHz. Besides meeting these specifications, the following problems have to be solved with respect to the high frequencies of operation: (a) feeding of the filters with low-loss half-wavelength transmission lines to improve the matching at the input and output ports; (b) providing high-quality transitions from the striplines embedded in the LTCC module to the coplanar ground-signal-ground test port for on-wafer probing. Because of the strict in-band requirements, the number of resonators constituting the filter could not be higher than four. In order to provide the required steepness of the filter skirts,two designs with transmission zeros at finite frequencies have been chosen: a coupled-line filter design and a cross-coupled filter design.A. Coupled-line Filter Design For the coupled-line filter, we applied the established design of bandpass filters based on half-wavelength coupled resonators with attenuation zeros 8. With this method, attenuationpoles are obtained by both input/output and inter-stage coupling.Tap-coupling,parallel-coupling and anti-parallel coupling structures were investigated. For the tap-coupling structure, an attenuation pole is generated at that frequency at which the electrical length from the phase centre of the tap to the open end of the resonator becomes 90. In order to generate an attenuation pole at pf0, a tap-coupling should be devised at a position where the electrical length tap from the open is equivalent to 90/p at centre frequency。 For the parallel-coupled lines structure, transmission becomes zero at the frequency where the electrical length of the coupled line becomes 180. Therefore, to generate an attenua-tion pole at pf0, the electrical length of the coupled line at centre frequency p has to be chosen 180/p . Since the electrical length of a half-wavelength resonator is 180, p will exceed 1. This implies that an attenuation pole can only be obtained at frequencies above the pass-band. For the anti-parallel coupled structure, where the open ends of the parallel lines are placed side by side, a pole is generated at pf0 under the condition ap = 90/p, where ap is the electrical length of the coupled line at centre frequency. As for the tap coupling, the attenuation pole can be obtained both below and above the pass-band. Two tap-couplings and one anti-parallel coupling have been used to design a four-pole bandpass filter with three attenuation zeros placed at 16.82 GHz, 22.43 GHz, and 32.88 GHz. The filter layout is shown in Fig. 1, panel (a). Half-wavelength feed-lines have been employed to reduce the influence of the feed-line impedance and provide ,therefore,good impedance matching. The initial simulation of the filter structure was performed by a 2.5-dim electromagnetic field simulator (AWR Microwave Office). Verification was carried out with a 3-dim full wave simulator (CST Microwave Studio). The simulated frequency response of the filter is presented in panel (b) of Fig. 1. The in-band insertion loss is less than 1.2 dB. The return loss is not worse than 14.5 dB. The characteristic slopes at the band edges amount to about 20 dB/GHz.图1 Four-pole coupled-lines bandpass filter with three attenuation poles: layout (a) and simulated frequency response (b).B. Cross-coupled Filter Design The synthesis of bandpass filters with source-load coupling was theoretically described and experimentally verified, especially for two-pole filters . Such a cross-coupling allows to obtain a frequency response with equal numbers of transmission poles and zeros. The same source-load coupling can be applied to higher-order filters as well, allowing for additional stop-band attenuation. The scheme of a four-pole cross-coupled filter is depicted in Fig. 2, panel (a). The resonators are represented by nodes, and the couplings are indicated as connecting lines. Two additional couplings have been added to the filter. The coupling C1 is a capacitive coupling between the input and the output feedlines of the filter; the coupling C2 is an inductive crosscoupling between the first and the fourth resonator. Adjusting the strength of the cross-coupling, the positions of the transmission zeros and the steepness of the filter slopes could be tuned to the desired values. The topology of the four-pole cross-coupled filter is shown in panel (b) of Fig. 2. The filter was implemented using LTCC multi-layer ceramics and consisted of four C-shaped stripline resonators, situated in two conductive layers: the first and the fourth resonator lines were printed in the bottom layer, while the second and the third resonator as well as the feed-lines were placed on the top conductive layer. The separation of the resonators in the vertical direction enables the reduction of the overall size of the filter, compared with an entirely planar structure. The numerical simulations were performed in two steps as described above. The expected frequency response of the cross-coupled filter is shown in Fig. 2, panel (c). Two pairs of transmission zeros, placed symmetrically around the pass-band, are clearly visible in both stop-bands of the filter. The in-band insertion loss IL was not worse than 1.4 dB, and the return loss RL better than 15 dB. The area occupied by the filter was 55 mm2. The steepness of the skirts amounted to 45 dB/GHz. Comparing the responses of the two types of filter (Fig. 1 and Fig. 2) reveals that the cross-coupled filter provides much higher attenuation in the narrow stop-band but, atfrequencies around 32-35 GHz, the performance suffers from a spurious harmonic, which was suppressed by one of the transmission zeros in the coupled-line filter design.图2 Four-pole cross-coupled filter: (a) schematic, (b) multi-layer layout, (c) simulated frequency response.III. FABRICATION AND MEASUREMENTS OF THE FILTERS Both filter structures were manufactured using the DuPont Green Tape 951 LTCC system with a thickness of the dielectric layers of 205 m (DP-951 AX/PX) and a nominal dielectric permittivity r = 7.8 after sintering This material system has been chosen for fabricaton, because it includes a variety of appropriate inks, photoimageable screen printing (Fodel-technology) 12, and provides satisfactory hermeticity and planarity of the sintered modules. Fig. 3 shows a photograph of LTCC-integrated Ka-band filters. 图3 LTCC module with the filter samples fabricated.Two different conductive materials were used for the fabrication: photo defined silver paste with a thickness of 7 m after sintering, and a laser-cut silver foil with a thickness of 25m. The built stack of LTCC-layers integrating the developed filters was laminated with a pressure of 20 MPa and a temperature of 70C for 10 minutes. The co-fired sintering was conducted in a muffle furnace using a firing profile with 875C peak temperature for 10 minutes and an overall duration of about 450 minutes. The electrical characterisation of the filters was conducted using an Agilent E8367A PNA vector network Analyser (up to 67 GHz) and 200-m-pitch coplanar ground-signal-ground probe tips (|Z|-probes from Suess MicroTec). The measured performance of the two different coupled-line filter samples is shown in Fig. 4. We observed a shift of the measured frequency response compared to the numerical expectation by about 500 MHz to higher frequencies. This effect could be attributed to a lower dielectric permittivity, r =7.5 as measured with a split-post dielectric resonator, compared to the value 7.8 used in the design. The in-band insertion loss was about 1 dB higher than expected from simulations, possibly due to ohmic losses in the ground planes, which were neglected in the simulation, and additional losses at the edges of the striplines due to significant roughness of both paste and foil metallisations. The return loss was better than 14 dB, in accordance with the simulations. There is apparently no difference in the insertion loss for the two versions employing silver paste and laser-cut foil metallisation.图4 Measured performance of two samples of the coupled-line filter: withsilver paste Fodel-metallisation (red), and laser-cut silver foil (blue).In order to avoid EMC problems and decrease mutual interactions with external circuitry, the filter structures were shielded by ground planes at the top and bottom layers(triplate-concept), and by surrounding via-fences connecting top- and bottom ground-layer along the contour Fig. 1, panel(a). The space between adjacent vias along the fence amounted to 315 m, a fired via diameter was about 130 mThe distance between the resonator line sections and the grounding vias constituting the fence was varied, in order to suppress the parasitic excitation of waveguide modes. Finally,two different nominal signal-to-ground distances were selected for fabrication: 1000 m and 600 m.According to Fig. 5 for the coupled-line filter, for the larger distance, waveguide modes were found to be excited already at frequencies around 28 GHz, while for the smaller distance this effect could be shifted to 35 GHz and, at the same time, the stop-band rejection was higher by more than 20 dB. On the other hand, the smaller distance caused slightly higher inband insertion loss due to stronger wall currents in the dissipative via-fence and ground planes.图5 Measured performance of the coupled-line filter for two signal-togrounddistances : 1000 m (blue) and 600 m (red).The measured performance of the cross-coupled filter is shown in Fig. 6 for 840 m and 525 m signal-to-ground distances, respectively. The different distances caused little differences in the frequency response. The in-band performance was practically identical to the measured coupled-line samples: The same shift of the frequency response to higher values is observed as for the previous case, due to the deviation between simulated and measured values of the dielectric permittivity. Also, the insertion loss was approximately 1 dB higher than simulated; the input matching was not worse than 12 dB. Four transmission zeros can be easily identified in the measured data, providing the desired steepness of the filter skirts and high out-of-band rejection.图6 Measured performance of the cross-coupled filter for two signal-togrounddistances: 840 m (blue) and 525 m (red).IV. CONCLUSIONS Two compact bandpass filters based on LTCC multi-layer technology were designed for potential applications in satellite communication downlinks at Ka-band frequencies. The fourpole bandpass filters with additional transmission zeros at finite frequencies use half-wavelength resonators and differ in the way how the transmission zeros are achieved. In one type, they are obtained by tap-coupling and anti-parallel coupling in the coupled-line structures. The other type employs cross-couplings between non-adjacent resonators as well as between the input and the output feed lines. The experimental investigation proved excellent agreement with numerical simulations and reasonable reproducibility of the manufacturing process. All four filters under test displayed good performance in terms of low in-band insertion loss, steep skirts, and high out-of-band rejection. Altogether, this work can be considered a further step towards the development of compact microwave modules, suitable for on-board signal processing for satellite communications. ACKNOWLEDGMENT This work was conducted in the framework of the project KERAMIS, which was funded by the German Aerospace Center (DLR) in charge of the German Federal Ministry of Economics (BMWi) under contract 50YB0313. The authors would like to gratefully acknowledge valuable scientific and technical contributions from D. Frster, M. Hintz, M. Huhn, and R. Perrone. Ka波段卫星的带通滤波器基于LTCC的通信应用 摘要为了Ka波段的卫星通信应用（下行17-22 GHz的）两个紧凑的带通滤波器的设计被提出。在有限的频率中设计两个有额外传输零点的滤波器，以提高带外选择性。过滤器在晶圆探测上已实现低成本LTCC模块和散射参数测量。一、 导言 现代多媒体卫星正变得日益复杂，如转发器连接要求，多次覆盖梁，等等。因此，卫星运营商们越来越多地要求灵活性功能，高效的信号路由和信号处理。因此，在多媒体卫星通信的未来的应用中，要求高的创新组件，RF性能，在同一时间，又需要低体重，小尺寸和高可靠性。先进的集成和封装技术，如低温共烧陶瓷（LTCC），结合不同设计技术，造价低，体积小，多功能。低温共烧陶瓷技术提供了模块化，混合集成系统具有高度的小型化微波有效载荷设备，因此，灵活性为适应不同的应用。为什么LTCC变得非常流行,这是这是原因之一，不仅因为低频设计，而且因为高频率的微波甚至毫米波范围。本文着重介绍Ka波段下行的带通滤波器多媒体卫星信号的重要组成部分的频率链，这是基于LTCC的多层技术，可以结合其他LTCC Ka波段微波目前正在开发的模块。为了Ka波段上行频率的有效抑制，和低插入损耗，这里介绍两个显示高的阻带隔离的过滤器。他们开发的分离通道组总带宽高达500 MHz到1 GHz，频率范围在17-22 GHz。出于演示目的，这些过滤器已发展为中心频率f0= 19.5 GHz。已花费额外的努力调查接地和不同的导电性的影响粘贴上过滤器性能的影响。二低温共烧陶瓷带通滤波器在有限频率的传输零点设计带通滤波器的设计已选定以下规格： -通带：18000 21000MHz； -最大带内插入损耗：2分贝； -最低带内回波损耗：12分贝（驻波比1.7）； -过滤器的斜坡的陡度：20分贝/GHz；除了满足这些要求，对于高频率操作还必须解决以下问题：(一)喂养低损耗半波长滤波器传输电路，以提高输入和输出端口的匹配；（二）提供高品质的过渡，从嵌入在LTCC模块共面带状线晶圆上探测地面-信号-地面测试端口。 因为严格的带内的要求，构成过滤器谐振器数量不能超过四个。为了提供所需的过滤器裙陡：在有限的频率上具传输零点的设计选择为：耦合线滤波器的设计和交叉耦合滤波器的设计。A. 耦合线滤波器设计 对于耦合线滤波器，我们采用了基于半波长耦合具有衰减零点8的谐振器设计带通滤波器。用此方法，衰减极点都得到了输入/输出级间耦合。抽头耦合，并行耦合和反平行耦合结构进行了研究。对于抽头耦合结构，在该频率产生衰减极点，电长度从谐振器抽头的中心相到开口端变为90 。为了在P F0产生一个衰减极点，抽头耦合应在从公开的长度tap是相当于90 /在中心P频率的位置上设计。 对于平行耦合线结构，传输在频率上变为零当耦合线电气长度变为180 。因此，要在P F0产生一个衰减极点，在中心频率p的耦合线电气长度必须选择180 / P。由于半波长谐振器的电长度为180 ，P将超过1。这意味着只能在通频带以上的频率获得的衰减极点。 对于反平行耦合结构，平行线的开路端并排放置，一极是产生在P F0=90 / p，其中ap耦合线电气长度的条件下，ap在中心频率。抽头耦合，可以得到以下及以上的通带衰减极点。 两个抽头耦合和一个反平行耦合已经被用于设计一个具有三个衰减零点的四极带通滤波器，三个衰减零点放置在16.82 GHz，22.43 GHz和32.88 GHz。该过滤器的布局如图1，面板（A）。已采用半波长馈线，以减少馈线的阻抗和供应的影响，因此，并提供良好的阻抗匹配。初始模拟滤波器结构是由2.5调光的电磁场仿真（AWR的微波办公室）。验证进行了三维全波仿真（CST微波工作室）。模拟频率响应的滤波器是在图1面板（B）上。带内插入损耗小于1.2分贝。回波损耗不超过14.5分贝差，在频带边缘量的特点斜坡约20 dB/ GHz。图2.1 带有3个衰减极点的四极耦合线带通滤波器：布局（a）和模拟的频率响应（b）项 B.交叉耦合滤波器设计 源负载耦合带通滤波器的合成，是经过理论阐述和实验验证的，尤其是两个极点滤波器。这种交叉耦合允许获得与传输极点和零点的数目相等的频率响应。相同的源负载耦合，可应用于高阶滤波器，以及允许额外的阻带衰减。 该计划的一个四极耦合滤波器，如图2，面板（A）所示。谐振器是由节点，和联轴器表示连接线。另外2个联轴器已被添加到过滤器。耦合C1是滤波器输入和输出间馈线的电容耦合；耦合C2是一个第一和第四谐振器之间的电感交叉耦合。交叉耦合C 1、C 2提供对称的双传输零点的频率响应。调整交叉耦合的强度，传输零点的位置，过滤器的斜坡的陡度可调谐到所需的值。 四极交叉耦合滤波器的拓扑结构如图2，显示板（B）所示。滤波器实现使用低温共烧多层陶瓷并由四个C形带状线谐振器组成，位于两个导电层：第一和第四谐振线印在底层，而第二和第三谐振器以及馈线放置在顶部的导电层。与完全平面结构相比，在垂直方向的分离谐振器，能够减少过滤器的整体规模。如上所述，在两个步骤进行了数值模拟。预期的交叉耦合滤波器的频率响应如图2，面板（C）所示。两双传输零点，对称放置在通带，在带阻滤波器中是清晰可见的。带内插入损耗IL不超过1.4分贝，而超过15分贝的回波