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西安文理学院本科毕业设计（论文）Soft Starting of Induction Motor With Torque ControlAdemir Nied, Member, IEEE, Jos de Oliveira, Rafael de Farias Campos,Rogrio Pinho Dias, and Luiz Carlos de Souza MarquesAbstractThis paper presents a simple technique based onstator-flux estimation to control the electromagnetic torque of in-duction motors (IMs) during soft starting. The inherent problemsrelated to pure integration of the back electromagnetic force toestimate the stator flux are minimized using the low-pass-fllterapproach. The experimental results are dealt with and comparedwith the usual current-control technique. The results obtained val-idate the proposed technique, showing its viability in applicationswhere the objective is to fit the IM torque profile during startingor stopping according to the load torque.Index TermsInduction motor (IM), soft starter, torquecontrol.I. I NTRODUCTIONTHE ac motor starters employing power semiconductorsare being increasingly used to replace electromagneticline starters and conventional reduced-voltage starters becauseof their controlled soft-starting capability with limited startingcurrent 1.Thyristor-based soft starters are cheap, simple, reliable, andoccupy less volume, and therefore, their use is a viable solutionto the induction motor (IM) starting problem 2.The angle-ramp technique is known as the voltage-ramptechnique because it does not have a voltage feedback appliedto the motor. It means that the IM voltage is controlled througha firing-angle ramp of the thyristors in an open loop. Thistechnique is simple, and it is used in low-cost commercial softstarters. It will always produce a starting quadratic torque curvethat can be applied to small hydraulic pumps and small fans.Depending on the initial switching instants of all the threephases to the supply, an IM may produce severe pulsationson the electromechanical torque, regardless of whether it iscontrolled by a direct-online starter or a soft starter 3.The electromagnetic-torque pulsations may cause shocksto the driven equipment and damage to mechanical systemcomponents, such as shafts, couplings, and gears, immediatelyPaper 2009-IDC-024.R3, presented at the Industry Applications SocietyAnnual Meeting, Edmonton, AB, Canada, October 59, and approved forpublication in the IEEE TRANSACTIONS ON INDUSTRY APPLICATIONS bythe Industrial Drives Committee of the IEEE Industry Applications Society.Manuscript submitted for review January 31, 2009 and released for publicationOctober 30, 2009. First published March 25, 2010; current version publishedMay 19, 2010. This work was supported by WEG Equipamentos Eltricos.A. Nied, J. de Oliveira, and R. de Farias Campos are with the De-partment of Electrical Engineering, Universidade do Estado de SantaCatarina, Joinville 89223-100, Brazil (e-mail: dee2anjoinville.udesc.br;dee2jojoinville.udesc.br; camposrafa.br).R. P. Dias is with WEG Equipamentos Eltricos, Jaragu do Sul 3000, Brazil(e-mail: rogeriopd).L. C. de Souza Marques is with the Power Electronics and Control ResearchGroup, Universidade Federal de Santa Maria, Santa Maria 97105-900, Brazil(e-mail: marques_lcsm).Digital Object Identifier 10.1109/TIA.2010.2045335if the strength of materials is exceeded, or in the long term,owing to fatigue 2.Numerous attempts have been made on the performanceanalysis and control techniques of a three-phase IM fed froma thyristorized voltage controller 49. In 8, a dynamicfunction was used for the thyristor triggering angle in thevoltage controller proving to be a simple and effective way toimprove transient performance. The rate at which the main fiuxbuilds up is decreased, and the transient torque is smoothed byemploying a proper triggering function.In 1, some control strategies are proposed to eliminateelectromagnetic-torque pulsations, both at starting and reclos-ing, and to keep the line current nearly constant at a presetvalue over the entire starting period. The proposed current-control strategy is composed of successive cosines and constantfunction segments of thyristor triggering angle.In 2, the performance of the IM during voltage-controlledsoft starting has been optimized by eliminating the supply-frequency torque pulsations using a pulsating-torque elimina-tion strategy applied in 10 and 11, by extending it to coverall of the operating conditions of a soft starter, and by keepingthe line current constant at the preset value over the entire soft-starting period.Thus, in 1 and 2, the technique used to start the IMwith soft starter is based on the close-loop control with currentlimitation.This current-control technique can provide a constant torqueduring the beginning and in the middle of the starting process.However, during the final period of this process, one can verifythe existence of a torque pulse, in which, for certain kinds ofloads, an abrupt acceleration can be generated. Therefore, itseems that a good solution to solve this problem is to directlycontrol the electromagnetic torque during the starting process.In this paper, a torque-control technique is proposed toeliminate the electromagnetic-torque pulsations and to keep theline current nearly constant at a preset value over the entire soft-starting period. This strategy allows the torque to be controlledin such a manner that a constant torque or even a tailored torquecan be followed by the IM during starting or stopping process.The proposed technique, besides making the benefitsobtained from the current-control technique previously citedpossible, also allows canceling the torque pulse and the accel-eration verified at the end of the IM starting.The control, protection, and monitoring functions of theproposed technique are implemented in a microcontroller. The-oretical results are verifled experimentally using a custom-designed experimental setup.This paper is organized as follows. Section II describes theproposed torque-control technique and operation principles;NIED et al.: SOFT STARTING OF INDUCTION MOTOR WITH TORQUE CONTROLFig. 1. Proposed control technique schematic diagram.Fig. 2. Proposed control technique detailed diagram.Section III shows the approach used to estimate the flux fromwhich the electromagnetic torque will be obtained; Section IVpresents experimental results of the proposed technique, and finally, conclusions are given in Section V.II. S YSTEM D ESCRIPTION AND O PERATION PRINCIPLESThe schematic diagram of the IM soft starter with the pro-posed torque-control technique is given in Fig. 1.The electromagnetic-torque estimation is obtained using alow-pass filter (LPF) as the stator-flux estimator which elim-inates the analog input offset. The closed-loop torque controlallows the production of the desired start-torque profile from theadopted reference torque (TREF), adjusting the electromagnetictorque (Tem) to the load torque.Fig. 2 shows a detailed diagram of the IM soft starter withthe proposed torque-control technique.This diagram is composed of three pairs of back-to- back-connected thyristors, a microcontroller-based control and protection circuitry, ring and analog interface circuits, synchronism, and humanmachine interface. The analog inter- face circuit receives the three line-to-line input voltage signals (VRS, VST, and VTR), the three line-to-line output voltage signals (Vab, Vbc, and Vca) via differential ampliers with resistive decades, and the three line current signals (ia , ib , and ic) via current transformers.The generation of thyristor firing pulses is obtained byusing three signals: the torque reference signal, the estimatedtorque signal, and the input-voltage synchronism signals. TABLE IDEFINITION OF OPERATION MODESFig. 3. Firing sequence at the beginning of starting process.Fig. 4. Firing sequence with three supply voltages (steady state).The estimated torque signal is compared with the torque referencesignal, generating an error signal. This error signal generatesa voltage signal at the proportionalintegral (PI) output. Thedeflnition of the thyristor triggering angle is made through apercent relation between the PI output and the maximum supplyvoltage value. That signal is converted into time, taking intoconsideration the electric supply system operation frequency.The time signal generated is used together with the input-voltage synchronism signal to generate a delay in each of thephases, which is proportional to the error signal. The instantthese signals cross zero is used to generate the SCRs firingpulses related to the respective phases.Therefore, as a function of the torque error signal, a variationof the firing angle is obtained which allows the control of thevoltage applied to the motor and, consequently, of the machinetorque.At any time, by implementing the thyristor controller, theIM operates in one of the operation modes defined in Table I:mode 0no supply voltage, mode 1, 2, and 3two supplyvoltages, and mode 4three supply voltages.Fig. 5. Phase voltage and current waveforms: (a) During starting process and (b) during steady state.Fig. 6. Diagram of the proposed electromagnetic-torque estimator based on an INT with offset minimization.Considering the thyristor firing angle , the train of trig-gering pulses required for the whole three-phase system willbe separated by an angle of 60. To allow variation in thevalue of with time, the train of the triggering pulses requiredwill be separated by 60 (t). In Fig. 3 the firing angle has a maximum value (120), i.e., minimum voltage, withunsymmetric two-phase operation modes.Fig. 4 shows the three-phase operation mode. The thyristorsare controlled in the same way as in modes 1, 2, and 3, but, in this case, it can be seen that the ring angle has a minimumvalue (60 ), i.e., maximum voltage.The IM supply voltage and current waveforms during start- ing process and steady state are shown in Fig. 5, where the angle ( ) has its maximum (a) and minimum value (b).For normal steady-state operation, with current continuouslyowing in the three phases, the triggering angle is = ,where is the load-dependent phase-shift angle. For , thethyristor conduction requirements are not met, and therefore,the constraint for triggering angle is .III. F LUX -E STIMATION A PPROACHThe IM stator flux is estimated through the integration ofthe back electromotive force as described by the followingequations in the - stationary reference frame 12, 13:where Vs and Is are the measured stator supply voltages andcurrents, respectively, and Rs is the stator resistance. Thismethod was chosen because it requires only one parameter, thestator resistance, which is obtained from known methods. The ux magnitude and angle of the estimated stator ux can be written as Thus, the IM torque can be calculated as follows:where p is the number of poles of the IM.When a pure integrator (INT) is implemented in the dis-crete form, as in a digital signal processor, an error canarise 12,13. This error comprises the drift produced bythe discrete INT and also the drift produced by measurementoffset error present in the back electromotive force. A smalldc component, no mater how small it is, can drive the pureINT into saturation. The integration error associated with theimplementation of the INT is constant and somehow looks likean offset in initial integrated value. From the signal at the inputof the INT, it is not easy to know whether the integrated signalwill have an offset or not.The measurement offset can be reduced to an accept-able value by introducing an offset adjustment at the inputof the INT 12, or this can be done using a high-pass lter (HPF) 15.The integration drift problem due to dc offset and measure-ment noise can be avoided using an LPF 13, 14.Thus, Fig. 6 shows the approach used to estimate the IMstator ux and the electromagnetic torque. The purpose of thesix HPFs used after the stator voltages and current analog readings is to eliminate the offset of the analog inputs, while the purpose of the rst LPF is to substitute the pure INT, avoid- ing the problems regarding the use of the INT as previously mentioned. The instantaneous torque input of the second LPF presents periodic oscillations, being the most signicant one in the supply frequency 16. To eliminate those oscillations, an LPF has been used. The second LPF was designed in the samemanner as the first LPF, generating the electromagnetic ltered torque curve (Temf ).The lter project was based on the following specicationsdened to the proposed stator-ux estimator.1) The motor stator-ux estimation must operate under afrequency range from 50 Hz 15% to 60 Hz +15% (42 to70 Hz) with 50- or 60-Hz motors; two, four, six, and eight poles; currents from 9 to 1400 A; and feeding voltages from 220 to 575 V.2) The response dynamics of the stator-ux estimation must be compatible with a semicycle of the supply-voltage sinusoidal signal, i.e., 10 ms for 50 Hz and 8.33 ms for 60 Hz.3) The calculation of the ux must be done within the semicycle of the supply-voltagesinusoidal signal.4) A sampling rate of 250 s must be used to carry out the estimation and control routines.Once the stator-ux-estimator specications are dened, thelter project was based on the use of rst-order Butterworth analog lters. At rst, the HPFs were designed with xed cutoff frequency, one decade below the excitation frequency. The ex-citation frequency was dened as the average of the excitation frequency variation rate, i.e., 55 Hz. The LPFs followed the same design criteria used for the HPFs.After that, the analog lters were discretized by using the bilinear transformation (or Tustin Method) with a sampling rate of 250 s. One can observe that the sampling rate results in a Nyquist frequency much higher than the operation frequency specied in item 1), resulting in very low distortion values for the discretized signals.The discretization of the HPFs resulted in the following transfer function:From (6), one can observe the existence of a zero on the unitary circle and a pole that depends on the value of , which, in turn, depends on the cutoff frequency. Equation (6) was implanted in the microcontroller. The discretization of the LPFs resulted in the following transfer function:Table II IM dateand has the same value as that of (8). In this case, one can also see the existence of a zero on the unitary circle and a pole that depends on the value of , which depends on the cutoff frequency. Equation (9) was implemented in the microcontroller.The correction factor (CF) takes into consideration the dif-ferences between the magnitude and phase values of the pure INT in relation to the LPF, i.e.,By transforming the values found from polar to rectangular,one will ndOnce the CFs are obtained, multiply them directly by the output signals s and s of the rst LPF to obtain the corrected components of the estimated stator ux, i.e.,In relation to the HPF, one can observe that the magnitude errors in the range of the dened frequencies are practically zero, and therefore, there is no need to correct them. In relation to the phase delays, they are eliminated through the use of the LPFs.IV. E XPERIMENTAL R ESULTSExperiments were carried out to verify the feasibility of the proposed torque-control technique. Two WEG standard Ims with different power ratings were tested. Table II shows the motor data in physical units.Fig. 7. Experimental setup for Motor 1. (a) View of hardware. (b) View of control system. (Courtesy of WEG Equipamentos Eltricos.)Fig. 8. Experimental setup for Motor 2. (a) View of hardware. (b) View of control system. (Courtesy of WEG Equipamentos Eltricos.)For Motor 1, the experimental setup (see Fig. 7) comprised a WEG soft-starter model SSW-06 with torque control, a Magtrol dynamometer provided with a controller and load-torque indi-cation of 28 Nm (maximum value), and an oscilloscope. It is also provided with an electromagnetic brake which imposes a load torque, a PID controller which can be adjusted to obtain the desired dynamic response, and two analog programmable outputs that were used to indicate the torque and speed (in revolutions per minute) obtained in an oscilloscope. The torque sensor used had an adjustable bandwidth from 5 kHz to 1 Hz and allows tuning of the torque signal frequency limitation. The motor was submitted to a supply frequency variation obtained by means of a frequency inverter that, in turn, fed the soft starter, allowing a frequency variation from 40 to 70 Hz. An LC line filter was connected to the inverter output to filter pulsewidth modulation (PWM) voltages under pure sinusoidal voltages. The experimental results presented in this paper were obtained under 220 V and 50 Hz.For Motor 2, due to the high current and power values, the experimental setup (see Fig. 8) comprised a soft starter with torque control and rated current of 365 A, a dc machine (300 kW) working as a dynamometer, and an oscilloscope. To test this motor, due to the impossibility of measuring the torque because of the lack of proper equipment, the motortorque was estimated by the soft starter through the ana- log output, the speed was obtained through a tachogenerator (010 V) and the current was obtained through the current probe with the bandwidth of 10 kHzall signals were obtained from the oscilloscope.The IM starting waveforms of speed, rms line current, and shaft and estimated torque versus time are shown in Figs. 914. For a comparative assessment with the technique proposed in this paper, Figs. 9 and 13 show starting performances using the current-limit technique with motors 1 and 2, respectively. In addition, it must be noted that, for all experiments shown in this paper, the load torque (TL) is constant in spite of the different values.As can be seen in Figs. 9 and 13, despite the fact that the line current is limited, there is a torque pulse, i.e., the torque is almost constant except at the final period of the starting process when the torque increases, at least, two times. This behavior is not desired, and it may damage the mechanical system components. The torque-control strategy proposed in this paper, however, eliminates the torque pulse (Figs. 1012) and reduces the start- ing time of the motorload combination, as shown in Fig. 14. For purposes of comparison, Figs. 9 and 10 show the soft- starting curves of Motor 1 using the current-limit technique and the torque-control technique, respectively. In both cases, the load torque and soft-starting period are the same. Considering the current-control strategy, when starting the motor, the thyristor trigger angle is controlled until the current Fig. 9. Starting performance of the current-limit technique for Motor 1rms line current limit of I = 2.2 IN and load torque of TL = 0.1 TN , where TN is the rated torque (CH1: speed, 636 r/min/div; CH2: rms line current, 10 A/div; CH3: shaft torque, 0.2 TN /div).Fig. 10. Starting performance of the proposed torque-control technique for Motor 1 with one point of referenceconstant motor torque of T = 0.2 TN and load torque of TL = 0.1 TN (CH1: speed, 636 r/min/div; CH2: rms line current, 10 A/div; CH3: shaft torque, 0.2 TN/div).Fig. 11. Starting performance of the proposed torque-control technique for Motor 1 with two points of referenceinitial motor torque of T = 0.16 TN , nal motor torque of T = 0.23 TN, and load torque of TL = 0.1 TN (CH1: speed, 636 r/min/div; CH2: rms line current, 10 A/div; CH3: shaft torque of 0.2 TN/div).Fig. 12. Starting performance of the proposed torque-control technique for Motor 1 with three points of referenceinitial motor torque of T = 0.05 TN , intermediate motor torque of T = 0.24 TN, final motor torque of T = 0.22 TN, and load torque of TL = 0.1 TN (CH1: speed, 636 r/min/div; CH2: rms line current, 10 A/div; CH3: shaft torque of 0.2 TN /div). Fig. 13. Starting performance of the current-limit technique for Motor 2rms line current limit of I = 2 IN and load torque of TL = 0.06 TN (CH1: estimated torque 0.05 TN/div; CH2: rms line current, 200 A/div; CH3: speed, 331 r/min/div).Fig. 14. Starting performance of the proposed torque control technique for Motor 2 with three points of referenceinitial motor torque of T = 0.1 TN , intermediate motor torque of T = 0.14 TN , nal motor torque of T = 0.1 TN , and load torque of TL = 0.06 TN (CH1: estimated torque 0.05 TN /div; CH2: rms line current, 200 A/div; CH3: speed, 331 r/min/div). limit is reached and remains under this limit until the motor reaches its rated speed, when the thyristor trigger angle is minimum, i.e., the same as that of the supply voltage. The current can thus be kept constant at a predefined value during the whole motor starting time. However, with regard to the torque at the motor shaft, the existence of a torque pulse close to nominal rotation is due to the thyristor trigger angle which becomes minimum, imposing a feeding voltage and sudden acceleration of the motor.Similar current behavior is verified for the torque-control strategy, which can be obtained by setting TREF to a constant value, i.e., one point of reference. However, in this case, there is no torque pulse, the speed is quite smooth, and an abrupt variation is no longer recognized. Observe that the control is acting directly on the torque profile which is different from the current-control case, where the control did not act directly on the torque profile.In Fig. 11, one can obtain different starting performances from the torque control technique, setting two points of refer- ence for TREF. In this case, the line current is not constant, and it follows the shaft-torque reference behavior. Therefore, one must set the reference torque ramp in such a manner that the maximum line current value does not exceed the line current limit.As a consequence of the torque reference, the speed curve presents a quadratic pattern. Thus, according to particular operating conditions, the torque prole can be adjusted and, consequently, the speed prole. It means that, using the pro- posed torque-control strategy, a good acceleration prole can be tailored by pulse-free torque over the entire starting period. The third condition is shown in Fig. 12. In this case, three points of reference for TREF were considered. For the first point, the initial torque was tailored as a torque ramp. After that, the torque was tailored to have approximately a constant value, i.e., points two and three have similar values. It can be seen, in this example, that the rms line current follows the shaft-torque reference behavior. In this case, too, one must set the reference torque ramp in such a manner that the maximum line current value is not exceeded considering the line current limit. In turn, the speed waveform, as a con- sequence of the tailored reference torque, presents a smooth behavior, particularly in the initial starting period. Finally, Figs. 13 and 14 show the starting results obtained from Motor 2 using two different soft-starting strategies. As mentioned earlier, the current-limit technique can be used to limit the motor line current, but the torque has an undesired pulse (Fig. 13).Using the proposed torque-control technique, the torque pulse is eliminated, and the starting time of the motor-load combination can be reduced (Fig. 14). In Fig. 14, three points of reference for TREF were considered. In this case, unlike the waveforms shown in Fig. 12, a torque ramp of two different shapes was tailored: In the first-half starting period, a rise torque ramp was defined, and after that, a descent torque ramp was defined. As a consequence, the speed waveform looks like an S with a smooth behavior at the beginning and at the end of the starting period. The motor rms line current follows the shaft- torque reference behavior.Fig. 15. Torque-estimation error due to stator resistance variation under two different temperatures: a) t = 25 C and t = 0 C. (b) t = 90 C and t = 65 C (CH1, CH25.09 Nm/div).From Figs. 912 one can observe a similar behavior: The torque plunges regardless of the control techniqueeither by current or torque limitation. A more detailed observation, how- ever, will show that this torque-plunge behavior takes place immediately after the motor acceleration stops; in other words, at the beginning of the steady state. The reason for this behavior of the torque is based on the change of the motor condition.At the moment right before the beginning of the steady state, the motor is accelerating, and when the steady state starts, the amount of electromagnetic-torque pulsations refiected to the shaft depends on the parameters of the mechanical subsystem 16. In addition, observe that in the torque-control technique, these plunges are considerably smaller because the acceleration is kept (almost) constant while the motor starts. Asfinal remarks, it is important take into account the follow- ing considerations.In frequency inverters, the stator resistance can be deter- mined by a test at dc level where the applied voltage-to-current ratio is measured 17. A dc current equal to the rated machine current is forced into the machine windings by applying a single voltage space vector, the amplitude of which is PWM controlled. However, this test cannot be used to obtain the stator resistance without changing the hardware of the standard soft starters.The method used, thus, gets the motor plate data to estimate the average stator resistance as function of the average motor losses under steady-state operation. The sequence of the calcu- lations is based on the IM performance calculations 18. Fig. 15 shows the infiuence of stator-resistance variation and of the power variation of extra losses in relation to the value used to calculate the average stator resistance, for a temperature variation of 65 C. Therefore, to decrease the torque-estimation errors at most, the following stator-resistance value was used for the estimator calculations: Rs = 1.1 Rs (calculated). Torque control is a limitation of the torque according to the reference imposed. Therefore, when reclosing the motor, the accelerating torque at that moment will establish the starting.For example, if the speed does not decrease too much, the voltage-interruption instant will just be a step in the speed ramp, and the results will be similar to those shown in Figs. 11, 12,All the equations used to determine the torque are based on a balanced system. Therefore, an unbalance system will cause a torque-indication error. This error must be taken into consideration when choosing the torque limit, which will have to be a value above the minimum necessary to start the load. Estimated fiux pulsations due to unbalanced voltages will notprovide equivalent torque pulsations because the entire system is extremely filtered.V. C ONCLUSIONA simple technique to control the IM electromagnetic torque during soft starting has been presented. Using this technique, the motor torque can be tailored according to the load torque, and the acceleration can be maintained constant over the entire starting period.The proposed strategy eliminates the shaft-torque pulsations during the starting process, and it can reduce the starting time of the motorload set. It is composed of the following. 1) An HPF to overcome the measurement offset from the input analog signal to an acceptable value.2) An LPF used in place of an INT to avoid integration drift problem. The magnitude and phase errors associated with the stator-fiux estimation are made up by using a simple compensator which is based on steady-state operations. 3) An LPF to filter the electromagnetic motor torque from the stator fiux estimation.The experimental results show the good performance of the proposed technique which was implemented in a commercial product without additional cost.R EFERENCES1 G. Zenginobus, I. Cadirci, M. Ermis, and C. 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Ertan, and M. Rahman, “A solid state direct-on line starter for medium voltage induction motors with minimized current and torque pulsations,” IEEE Trans. Energy Convers., vol. 14, no. 3, pp. 402412, Sep. 1999.12 X. Xu and D. W. Novotny, “Implementation of direct stator fiux orienta- tion control on a versatile DSP-based system,” IEEE Trans. Ind. Appl., vol. 27, no. 4, pp. 694700, Jul./Aug. 1991.13 N. R. N. Idris, A. H. M Yatim, and N. A. Azli, “Direct torque control of induction machines with constant switching frequency and improvedstator fiux estimation,” in Proc. IEEE 27th Annu. Ind. Electron. Soc. Conf.,Denver, CO, 2001, pp. 12851291.14 D. Seyoum, F. Rahman, and C. Grantham, “Simplified fiux estimation for control application in induction machines,” in Proc. IEEE IEMDC, Jun. 2003, vol. 2, pp. 691695.15 L. A. Mihalache, “A fiux estimator for induction motor drives based on digital EMF integration with pre- and post-high pass filtering,” in Proc. IEEE Appl. Electron. Conf. 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New York: McGraw-Hill, 2003.Ademir Nied (M08) received the B.E. degree in electrical engineering from the Federal University of Santa Maria, Santa Maria, Brazil, in 1987, the M.S. degree in industrial informatics from the Fed- eral Technological University of Parana, Curitiba, Brazil, in 1995, and the Ph.D. degree in electrical engineering from the Federal University of Minas Gerais, Belo Horizonte, Brazil, in 2007.He is currently an Associate Professor in the De- partment of Electrical Engineering, State University of Santa Catarina, Joinville, Brazil. His current re- search interests are neural networks, intelligent control, and control of electrical drives.Jos de Oliveira received the B.E. degree in electri- cal engineering from the State University of Santa Catarina, Joinville, Brazil, in 1986, and the M.E. and Ph.D. degrees in electrical engineering from the Federal University of Santa Catarina, Florianpolis, Brazil, in 1994 and 2000, respectively. He is currently an Associate Professor in the De- partment of Electrical Engineering, State University of Santa Catarina. His current research interests are control applied to power electronics and electrical drives.Rafael de Farias Campos received the B.E. and M.E. degrees in electrical engineering from the State University of Santa Catarina, Joinville, Brazil, in 2001 and 2008, respectively.Currently, he is with the Department of Electrical Engineering, State University of Santa Catarina. His area of research is microprocessor control of motor drives.Luiz Carlos de Souza Marques received the B.E. degree in electrical engineering from the Federal University of Santa Maria, Santa Maria, Brazil, in 1981, the M.E. degree in electrical engineering from the Federal University of Santa Catarina, Florianpo- lis, Brazil, in 1996, and the Ph.D. degree in electrical engineering jointly from the Federal University of Santa Catarina and the Universit de Nantes, Nantes,France, in 2001.He is currently a Professor in the Department of Electromechanical and Power Systems, Federal University of Santa Maria. His current research interest is control applied to electrical motor drives.Rogrio Pinho Dias received the B.E. degree in electrical engineering from the Regional Univer- sity of Blumenau, Blumenau, Brazil, in 1998, and the M.E. degree in electrical engineering from the State University of Santa Catarina, Joinville, Brazil, in 2008.Currently, he is with WEG Equipamentos Eltri- cos, Jaragu do Sul, Brazil, where he is a Research and Development Engineer. His areas of research are induction motor soft starters and microprocessor control of motor drives.译文：异步电动机软启动与转矩控制摘要：本文提出了基于定子磁链估计的一个简单的方法来控制在软启动过程中异步电动（IMS）的电磁转矩。背面的电磁力来估计定子磁通的纯集成的固有问题就是指尽量减少使用低通滤波器的方法，这种方法是与背面的用来估计定子磁通的电磁力相关。实验结果被处理，并且会与通常的电流控制技术做比较。得到的结果验证所提出的方法，在应用程序中显示其可行性，目标是适合启动过程中的IM转矩曲线或根据负载转矩的停止。Keyword : 指数条款异步电动机（IM），软启动，转矩控制。一 导言采用功率半导体交流电动机启动器正在越来越多地用于取代电磁启动器和传统的降低电压启动，因为他们用有限启动电流1来控制软启动能力。基于晶闸管软启动器便宜，简单，可靠，体积小，此外，他们用一个可行的解决方案来解决异步电动机（IM）的启动问题2。角斜技术被称为斜坡电压技术，因为它并不适用于电机的电压反馈。这意味着，IM电压是通过在开环的晶闸管点火角的斜坡来控制的。这种技术很简单，它用于低成本的商业软启动器中。 它总是会产生一个开始的二次扭矩曲线，可应用于小型液压泵和小型电扇。所有三个阶段的初始开关瞬间取决于供应，机电扭矩的IM可能会产生严重的脉动，无论它是由直接在线启动或软启动器3控制。如果超过材料强度，或长期处于疲劳条件下2 ，电磁转矩脉动可能会导致驱动设备和机械系统部件损坏的冲击， 如轴，联轴器，齿轮。已多次尝试的性能分析和来自美联储thyristorized电压控制器4 - 9的控制技术。在8中，一个充满活力的功能，用于电压控制器中的晶闸管的触发角，证明是改善瞬态性能简单而有效的方式，主磁通建立率下降，采用适当的触发功能使瞬时转矩平滑。在1中，提出了一些控制策略，以消除在启动和重合闸过程中电磁转矩脉动，并保持该行目前在整个开始期间的预设值几乎不变。拟议的电流控制策略由连续余弦和晶闸管触发角的常数函数段组成。 在电压控制的软启动性能的IM已经得到了优化，这种优化是通过使用应用于10和11的脉动转矩消除策略来消除电源频率转矩脉动，通过扩大到包括所有软启动器的操作条件和保持整个软启动期间的预设值在并线的电流恒定范围内。因此，在1和2中，用于启动软启动器的IM技术 基于闭环控制电流限制。这个电流控制技术在开始和在启动过程中可以提供一个恒定的转矩. 然而，在这一进程的最后期限可以验证转矩脉动的存在，其中， 某些种类的负载，可以产生一个突然加速。因此，在启动过程中直接控制的电磁转矩似乎是解决这个问题的好办法。在本文中， 转矩控制技术提出了消除电磁转矩脉动和使线电流在整个软启动期间以一定的预设值保持不变。这个策略允许转矩以这样的方式来控制为恒定转矩，甚至可以遵循IM在启动或停止的过程中量身定制的扭矩所提出的方法，除了前面提到的从目前控制技术中获得的好处外，也可以取消转矩脉冲，并在年底的IM启动器中得到验证加速。拟议技术的控制，保护和监督职能作用在单片机中。使用定制设计实验装置对理论结果进行了验证。; 本文安排如下：第二节介绍了拟议的扭矩控制技术和操作原则；第三节显示用来估计从电磁转矩将获得通量的方法；第四节提出建议技术的实验结果；最后，结论是在第五。 图1 提出控制技术示意图图2 提出控制技术的详细图二 、系统操作与描述原则软启动器的IM提出的扭矩控制技术的示意图如图1通过使用一个低通滤波器（LPF）得到的电磁转矩估计作为定子磁链的估计，从而能消除模拟输入失调。闭环转矩控制，可以通过生产所需的启动转矩曲线作为参考转矩（TREF），来调整负载转矩的电磁转矩（TEM）。图2显示了一个建议的扭矩控制技术的IM软启动器的详细图。此图是由三对背靠背连接的晶闸管，基于微控制器的控制，保护电路，发射，模拟接口电路，同步，人机界面组成。模拟接口电路接收三对线对线输入电压信号（VRS时，VST，录像机），和通过电阻十年的差分放大器的三对线对线输出电压信号（VAB，VBC，VCA），和通过电流互感器（IA，IB，IC）的三线电流信号。晶闸管触发脉冲的生成是通过使用三个信号：转矩给定信号，估计扭矩信号，输入电压同步信号。估计扭矩信号与转矩给定信号相比较产生一个错误的信号，这种错误信号产生一个比例积分（PI）的输出电压信号。晶闸管的触发角的定义是通过 PI输出和最大供电电压值之间的百分比关系确定的。考虑到供电系统的运行频率，该信号被转换成时间。 Time(s)图3 启动过程的开始射击序列Time（s）图4 射击三个电源电压（稳态）序列产生的时间信号与输入电压同步信号一起使用，在每个阶段产生的延迟，是成正比的误差信号。这些信号的瞬间跨零被用来产生与晶闸管发射有关的各阶段的脉冲。因此，作为转矩误差信号的功能，得到射击角度的变化允许电压控制适用于电机和机器的扭矩控制中。在任何时间，通过实施晶闸管控制器，IM运行在一个操作模式之下定义了表1，模式0没有电源电压， 模式1，2，3有2个电源电压，和模式4有3个电源电图5 相电压和电流图形 （a）启动过程及（b）在稳定状态图6 基于偏移最小化诠释基础上的建议电磁转矩估计考虑晶闸管的触发角，整个三相系统所需的触发脉冲的阵列将相隔60角。允许在值随时间的变化过程中，所需的触发脉冲的阵列将被60（T）分离。在图3中，发射角有一个最大值（120），即最低电压不对称的两阶段的运作模式。图4显示了三个阶段的操作模式。晶闸管以同样的方式控制模式1，2和3，但是，在这种情况下可以看出发射角有一个最低值（60），即最大电压。图5显示了IM电源电压和电流波形在启动过程和稳态的波形。其中，角（ - ），有最大值（a）和最低值（b）。对于正常的稳态运行，目前不断流动中的三个阶段，触发角为=，其中是依赖负载的移相角。对于，可控硅导通的要求得不到满足，因此，触发角的约束是。三、 通量估计方法据估计，通过一体化的反电动势的IM定子磁通在-静止坐标系中用方程12 和13来描述：其中Vs和Is分别是测量定子供电电压和电流，RS是定子电阻。选择这种方法的原因是它要求只有一个参数，定子电阻可以从已知的方法获得。流量的大小和估计定子磁链的角度可以写成：因此，扭矩可以计算如下：其中p是IM的极数当一个纯积分器（INT）在离散的形式得以实现，在数字信号处理器中，一个错误可能会出现1213。此错误包括离散诠释产生的漂移和目前在反电动势中通过测量产生的偏移漂移误差。 一个小的直流分量，无论它多么小，都可以进入纯粹诠释的饱和状态。与INT实施相关的整合错误是个常数，并以某种方式看起来就像是抵消最初的综合价值。对于输入的INT信号而言，是不容易知道是否会有偏移或不集成信号的。 测量偏移量可以在INT12的输入端通过引入偏移调整可以减少到一个可接受的值，或可以做到使用高通滤波器（HPF）15。由于直流偏移和测量噪声的集成漂移问题，可避免使用低通滤波器1314。因此，图6显示了用于估计IM定子磁链和电磁转矩的方法。定子电压和电流模拟读数后的六个住房公积金的用途是消除模拟输入的偏移，而第一低通滤波器的目的是要取代纯粹的诠释，避免以前提及的在使用INT是所遇到的问题。第二个LPF的瞬时扭矩输出，在电源频率中是最重要的一个周期振荡16。为了消除这些振荡，低通滤波器已被使用。以同样的方式设计的 第二个低通滤波器作为第一LPF，产生过滤的电磁转矩曲线（Temf）。基于以下规格的过滤器项目定义了建议定子磁链的估计。1）电机定子磁链的估计，必须用50 - 或60-赫兹的电机在频率从50 Hz-15到60赫兹+15（42-70赫兹）的范围下运行，;二，四，六，八杆，电流从9A至1400A，喂养电压是从220到575 V。2）定子磁链的估计反应动力学必须与电源电压的正弦信号semicycle兼容，也就是说，10毫秒为50赫兹，8.33毫秒为60赫兹。3）通量的计算必须在semicycle内的电源电压的正弦信号中完成。4）必须使用250s的采样率进行估计和控制程序。 一旦定子磁链估计规格被定义，过滤器项目就是基于第一阶巴特沃斯模拟滤波器的使用。首先，住房公积金是用固定截止频率来设计的，下一个十年低于这个激励频率。激励频率的平均变化率称为激励频率，即，55赫兹。LPFs是遵循相同设计标准的住房公积金来使用的。在这之后，模拟滤波器离散双线性变换（或塔斯廷方法）用了250s的采样
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